Method of utilizing and manipulating wireless resources for efficient and effective wireless communication

ABSTRACT

A method of allocating symbols in a wireless communication system is disclosed. More specifically, the method includes receiving at least one data stream from at least one user, grouping the at least one data streams into at least one group, wherein each group is comprised of at least one data stream, preceding each group of data streams in multiple stages, and allocating the precoded symbols.

This application claims the benefit of U.S. Provisional Application No.60/801,689 filed on May 19, 2006, U.S. Provisional Application No.60/896,831 filed on Mar. 23, 2007, U.S. Provisional Application No.60/909,906 filed on Apr. 3, 2007, and U.S. Provisional Application No.60/910,420 filed on Apr. 5, 2007, which are hereby incorporated byreference.

BACKGROUND OF THE INVENTION

1. Field of the Invention

The present invention relates to a method of using wireless resources,and more particularly, to a method of utilizing and manipulatingwireless resources for efficient and effective wireless communication.

2. Discussion of the Related Art

In the world of cellular telecommunications, those skilled in the artoften use the terms 1G, 2G, and 3G. The terms refer to the generation ofthe cellular technology used. 1G refers to the first generation, 2G tothe second generation, and 36 to the third generation.

1G refers to the analog phone system, known as an AMPS (Advanced MobilePhone Service) phone systems. 2G is commonly used to refer to thedigital cellular systems that are prevalent throughout the world, andinclude CDMAOne, Global System for Mobile communications (OSM), and TimeDivision Multiple Access (TDMA). 2G systems can support a greater numberof users in a dense area than can 1G systems.

3G commonly refers to the digital cellular systems currently beingdeployed. These 3 G communication systems are conceptually similar toeach other with some significant differences.

In a wireless communication system, an effective transmission of datacrucial and at the same time, it is important to improve transmissionefficiency. To this end, it is important that more efficient ways oftransmitting and receiving data are developed.

SUMMARY OF TUE INVENTION

Accordingly, the present invention is directed to a method of utilizingand manipulating wireless resources for efficient and effective wirelesscommunication that substantially obviates one or more problems due tolimitations and disadvantages of the related art.

An object of the present invention is to provide a method allocatingsymbols in a wireless communication system.

Another object of the present invention is to provide a method ofperforming hierarchical modulation signal constellation in a wirelesscommunication system.

A further object of the present invention is to provide a method oftransmitting more than one signal in a wireless communication system.

Additional advantages, objects, and features of the invention will beset forth in part in the description which follows and in part willbecome apparent to those having ordinary skill in the art uponexamination of the following or may be learned from practice of theinvention. The objectives and other advantages of the invention may berealized and attained by the structure particularly pointed out in thewritten description and claims hereof as well as the appended drawings.

To achieve these objects and other advantages and in accordance with thepurpose of the invention, as embodied and broadly described herein, amethod of allocating symbols in a wireless communication system includesreceiving at least one data stream from at least one user, grouping theat least one data streams into at least one group, wherein each group iscomprised of at least one data stream, precoding each group of datastreams in multiple stages, and allocating the precoded symbols.

In another aspect of the present invention, a method of performinghierarchical modulation signal constellation in a wireless communicationsystem includes allocating multiple symbols according to abits-to-symbol mapping rule representing different signal constellationpoints with different bits, wherein the mapping rule represents one (1)or less bit difference between closest two symbols.

In a further aspect of the present invention, a method of transmittingmore than one signal in a wireless communication system includesallocating multiple symbols to a first signal constellation and to asecond constellation, wherein the first signal constellation refers tobase layer signals and the second signal constellation refers toenhancement layer signals, modulating the multiple symbols of the firstsignal constellation and the second signal constellation, andtransmitting the modulated symbols.

It is to be understood that both the foregoing general description andthe following detailed description of the present invention areexemplary and explanatory and are intended to provide furtherexplanation of the invention as claimed.

BRIEF DESCRIPTION OF THE DRAWINGS

The accompanying drawings, which are included to provide a furtherunderstanding of the invention and are incorporated in and constitute apart of this application, illustrate embodiment(s) of the invention andtogether with the description serve to explain the principle of theinvention. In the drawings;

FIG. 1 is an exemplary diagram of a generalized MC-CDM structure;

FIG. 2 is another exemplary diagram of a generalized MC-CDM structure;

FIG. 3 is an exemplary diagram illustrating a generalized MC-CDMstructure in which precoding/rotation is performed on groups;

FIG. 4 is an exemplary diagram illustrating a multi-stage rotation;

FIG. 5 is another exemplary diagram of a generalized MC-CDM structure;

FIG. 6 is an exemplary diagram illustrating frequency-domain interlacedMC-CDM;

FIG. 7 is an exemplary diagram illustrating an example of Gray coding;

FIG. 8 is an exemplary diagram illustrating mapping for regularQPSK/QPSK hierarchical modulation or 16 QAM modulation;

FIG. 9 is an exemplary diagram illustrating bits-to-symbol mapping for16 QAM/QPSK;

FIG. 10 is another exemplary diagram illustrating bits-to-symbol mappingfor 16 QAM/QPSK;

FIG. 11 is another exemplary diagram illustrating bits-to-symbol mappingfor 16 QAM/QPSK;

FIG. 12 is another exemplary diagram illustrating bits-to-symbol mappingfor 16 QAM/QPSK;

FIG. 13 is an exemplary diagram illustrating bits-to-symbol mapping forQPSK/QPSK;

FIG. 14 is an exemplary diagram illustrating an enhancement layerbits-to-symbol for base layer 0x0;

FIG. 15 is an exemplary diagram illustrating an enhancement layerbits-to-symbol for base layer 0x1;

FIG. 16 is an exemplary diagram showing the signal constellation of thelayered modulator with respect to QPSK/QPSK hierarchical modulation;

FIG. 17 is an exemplary diagram illustrating the signal constellation ofthe layered modulator with respect to 16 QAM/QPSK hierarchicalmodulation;

FIG. 18 is an exemplary diagram showing the signal constellation for thelayered modulator with QPSK/QPSK hierarchical modulation;

FIG. 19 is an exemplary diagram illustrating the signal constellation ofthe layered modulator with respect to 16 QAM/QPSK hierarchicalmodulation;

FIG. 20 is an exemplary diagram illustrating signal constellation forlayered modulation with QPSK base layer and QPSK enhancement layer;

FIG. 21 is an exemplary diagram illustrating the signal constellation ofthe layered modulator with respect to 16 QAM/QPSK hierarchicalmodulation;

FIG. 22 is an exemplary diagram illustrating Gray mapping for rotatedQPSK/QPSK hierarchical modulation;

FIG. 23 is an exemplary diagram illustrating an enhanced QPSK/QPSKhierarchical modulation;

FIG. 24 is an exemplary diagram illustrating a new QPSK/QPSKhierarchical modulation;

FIG. 25 is another exemplary diagram illustrating a new QPSK/QPSKhierarchical modulation; and

FIG. 26 is an exemplary diagram illustrating a new bit-to-symbol block.

DETAILED DESCRIPTION OF THE INVENTION

Reference will now be made in detail to the preferred embodiments of thepresent invention, examples of which are illustrated in the accompanyingdrawings. Wherever possible, the same reference numbers will be usedthroughout the drawings to refer to the same or like parts.

An orthogonal frequency division multiplexing (OFDM) is a digitalmulti-carrier modulation scheme, which uses a large number ofclosely-spaced orthogonal sub-carriers. Each sub-carrier is usuallymodulated with a modulation scheme (e.g., quadrature phase shift keying(QPSK)) at a low symbol rate while maintaining data rates similar toconventional single-carrier modulation schemes in the same bandwidth.

The OFDM originally does not have frequency diversity effect, but it canobtain frequency diversity effect by use of forward error correction(FEC) even in a distributed mode. That is, the frequency diversityeffect becomes low when the channel coding rate is high.

In view of this, multi-carrier code division multiplexing (MC-CDM) or amulti-carrier code division multiple access (MC-CDMA) with advancedreceiver can be used to compensate for low frequency diversity effectdue to high channel coding rate.

The MC-CDM or MC-CDMA is a multiple access scheme used in OFDM-basedsystem, allowing the system to support multiple users at the same time.In other words, the data can be spread over a much wider bandwidth thanthe data rate, a signal-to-noise and interference ratio can beminimized.

For example, with respect to signal processing, a channel response foreach OFDM tone (or signal or sub-carrier) can be modeled as identicalindependent complex Gaussian variable. By doing so and using MC-CDM,diversity gain and processing gain can be attained. I-ere, interference,such as inter-symbol interference (ISI) or multiples access interference(MAI), is temporarily omitted in part due to the cyclic prefix or zeropadding employed by OFDM or MC-CDM.

FIG. 1 is an exemplary diagram of a generalized MC-CDM structure.Referring to FIG. 1, $\overset{\sim}{H} = \begin{bmatrix}{\overset{\sim}{h}}_{1} & \quad \\\quad & {\overset{\sim}{h}}_{2}\end{bmatrix}$denotes the frequency response of fading channel, where {tilde over(h)}₁ is a complex Gaussian variable for the frequency-domain channelresponse of each sub-carrier. Furthermore, without loss of thegenerality, $U_{2} = \begin{bmatrix}\alpha & \beta \\{- \beta^{*}} & \alpha^{*}\end{bmatrix}$denote the unitary symbol preceding matrix with power constraint|α|²+|β|²=1. It can be taken a generalization of the classic MC-CDM.

The processes of FIG. 1 include channel coding followed by spreading andmultiplexing (which can be represented by U). Thereafter, themultiplexed data is modulated by using the OFDM modulation scheme.

At the receiving end, the OFDM modulated symbols are demodulated usingOFDM demodulation scheme. They are then despread and detected, followedby channel decoding.

Further to the generalized MC-CDM structure, other structures areavailable such as rotated MC-CDM, OFDM, rotational OFDM (R-OFDM), orWalsh-Hadamard MC-CDM.

With respect to rotated MC-CDM, if α=cos(θ₁) and β=sin(θ₁), then areal-value rotation matrix can be available as follows in Equation 1.$\begin{matrix}{{{R_{2}\left( \theta_{1} \right)} = \begin{bmatrix}{\cos\left( \theta_{1} \right)} & {\sin\left( \theta_{1} \right)} \\{- {\sin\left( \theta_{1} \right)}} & {\cos\left( \theta_{1} \right)}\end{bmatrix}}{{R_{2}^{- 1}\left( \theta_{1} \right)} = {{R_{2}^{H}\left( \theta_{1} \right)} = \begin{bmatrix}{\cos\left( \theta_{1} \right)} & {- {\sin\left( \theta_{1} \right)}} \\{\sin\left( \theta_{1} \right)} & {\cos\left( \theta_{1} \right)}\end{bmatrix}}}} & \left\lbrack {{Equation}\quad 1} \right\rbrack\end{matrix}$

Furthermore, with respect to OFDM, if αβ=0 or αβ*=0, then U, becomes I₂.In other words, U₂ becomes uncoded OFDM or uncoded OFDMA. In addition,with respect to Walsh-Hadamard MC-CDM, if${\alpha = {{\cos\left( \frac{\pi}{4} \right)} = {{\frac{\sqrt{2}}{2}{and}{\quad\quad}\beta} = {{\sin\left( \frac{\pi}{4} \right)} = \frac{\sqrt{2}}{2}}}}},$U₂=R₂ become a classic Walsh-Hadamard matrix.

FIG. 2 is another exemplary diagram of a generalized MC-CDM structure.In FIG. 2, a plurality of data are inputted which are then precodedand/or rotated. Here, the preceding or rotation also can signifyadjustment of the amplitude and/or phase of incoming data.

With respect to precoding/rotation, different tones or sub-carriers maybe precoded/rotated independently or jointly. Here, the jointprecoding/rotation of the incoming data or data streams can be performedby using a single rotation matrix. Alternatively, different incomingdata or data streams can be separated into multiple groups, where eachgroup of data streams can be precoded/rotated independently or jointly.

FIG. 3 is an exemplary diagram illustrating a generalized MC-CDMstructure in which precoding/rotation is performed on groups. Referringto FIG. 3, multiple data or data streams are grouped into Data Stream(s)1, 2, . . . , K groups which are then precoded/rotated per group. Here,the precoding/rotation can include amplitude and/or phase adjustment, ifnecessary. Thereafter, the precoded/rotated symbols are mapped.

Further, different rotation/precoding on different groups may lead to amixture of OFDM, MC-CDM or R-OFDM. In addition, the rotation/precodingof each group may be based on the QoS requirement, the receiver profile,and/or the channel condition.

Alternatively, instead of using a big precoding/rotation matrix, asmaller-sized precoding/rotation matrix can be dependently orindependently applied to different groups of incoming data streams.

In operation, actual precoding/rotation operation can be performed inmultiple stages. FIG. 4 is an exemplary diagram illustrating amulti-stage rotation. Referring to FIG. 4, multiple data or data streamsare inputted which are then precoded/rotated. Here, these processedsymbols can be grouped into at least two groups. Each group isrepresented by at least one symbol.

With respect to rotation of the symbols, the symbol(s) of each group canbe spread using a spreading matrix. Here, the spreading matrix that isapplied to a group may be different and can be configured. After thesymbols are processed through the spreading matrix, then the output(s)can be re-grouped into at least two groups. Here, the re-grouped outputscomprise at least one selected output from each of the at least twogroups.

Thereafter, these re-grouped outputs can be spread again using thespreading matrix. Again, the spreading matrix that is applied to a groupmay be different and can be configured. After the outputs are processedthrough another spreading matrix, they are inputted to an inverse fastFourier transform (IFFT).

A rotation scheme such as the multi-stage rotation can also be employedby a generalized MC-CDM or multi-carrier code division multiple access(MC-CDMA). FIG. 5 is an exemplary diagram illustrating a general blockof the MC-CDM.

FIG. 5 is another exemplary diagram of a generalized MC-CDM structure.More specifically, the processes as described with respect to FIG. 5 aresimilar to those of FIG. 1 except that FIG. 5 is based on generalizedMC-CDM or MC-CDMA that uses rotation (e.g., multi-stage rotation). Here,after channel coding, the coded data are rotated and/or multiplexed,followed by modulation using inverse discrete Fourier transform (IDFT)or IFFT.

At the receiving end, the modulated symbols are demodulated usingdiscrete Fourier transform (DFT) or fast Fourier transform (FFT). Theyare then despread and detected, followed by channel decoding.

In addition, interlacing is available in the generalized MC-CDM. In 1xevolution data optimized (1xEV-DO) BCMCS and enhanced BCMCS (EBCMCS),the multipath delay spread is about T_(d)=3.7 μs and the coherentbandwidth is around $B_{c} = {\frac{1}{T_{d}} = {270{{kHz}.}}}$Therefore, the maximum frequency diversity order is$d = {\frac{B}{B_{c}} = {\frac{1.25}{0.27} \approx 5.}}$This means, in order to capture the maximum frequency diversity here,the MC-CDM spreading gain L≧5 is possibly enough.

Based on the above analysis, a frequency-domain interlaced MC-CDM can beused. FIG. 6 is an exemplary diagram illustrating frequency-domaininterlaced MC-CDM. Referring to FIG. 6, each slot, indicated bydifferent fills, can be one tone (or sub-carrier) or multipleconsecutive tones (or sub-carriers).

The tone(s) or sub-carrier(s) or symbol(s) can be rotated differently.In other words, the product distance, which can be defined as theproduct of Euclidean distances, can be maximized. In detail, a minimumproduct distance, which is used for optimizing modulation diversity, canbe shown by the following equation. The minimum product distance canalso be referred to as Euclidean distance minimization. $\begin{matrix}{D_{p} = {\min{\prod\limits_{{i \neq j},{s_{i} \in A}}{{s_{i} - s_{j}}}}}} & \left\lbrack {{Equation}\quad 3} \right\rbrack\end{matrix}$

Referring to Equation 3, s_(i)εA denotes the transmitted symbols.Furthermore, optimization with maximizing the minimum productiondistance can be done by solving the following equation. $\begin{matrix}{{U_{2}\left( {\mathbb{e}}^{j\phi} \right)} = {{\underset{U}{\arg\quad\max}D_{p}} = {\underset{U}{\arg\quad\max}\quad\min{\prod\limits_{{i \neq j},{s_{i} \in A}}{{{Us}_{i} - {Us}_{j}}}}}}} & \left\lbrack {{Equation}\quad 4} \right\rbrack\end{matrix}$

Referring to Equation 4,${U_{2}\left( {\mathbb{e}}^{j\phi} \right)} = {\begin{bmatrix}\alpha & {\alpha\mathbb{e}}^{j\phi} \\{{- \alpha^{*}}{\mathbb{e}}^{- {j\phi}}} & \alpha^{*}\end{bmatrix}.}$

For example, for the traditional quadrature phase shift keying (QPSK),U₂(e^(jφ)) can be decided by calculating${d\left( {\mathbb{e}}^{j\phi} \right)} = {\frac{1}{2}{{\Delta_{1}^{2} - \left( {{\mathbb{e}}^{j\phi}\Delta_{2}} \right)^{2}}}}$where Δ_(1,2)ε{±1, ±j, ±1±j}.

As discussed, each tone or symbol can be rotated differently. Forexample, a first symbol can be applied QPSK, a second symbol can beapplied a binary phase shift keying (BPSK), and n^(th) symbol can beapplied 16 quadrature amplitude modulation (16 QAM). To put differently,each tone or symbol has different modulation angle.

In rotation OFDM/MC-CDM (R-OFDM/MC-CDM),$\hat{H} = {{\overset{\sim}{H}U_{2}} = {{\begin{bmatrix}{\overset{\sim}{h}}_{1} & \quad \\\quad & {\overset{\sim}{h}}_{2}\end{bmatrix}\begin{bmatrix}\alpha & \beta \\{- \beta^{*}} & \alpha^{*}\end{bmatrix}} = {\begin{bmatrix}{{\overset{\sim}{h}}_{1}\alpha} & {{\overset{\sim}{h}}_{1}\beta} \\{{- {\overset{\sim}{h}}_{2}}\beta^{*}} & {{\overset{\sim}{h}}_{2}\alpha^{*}}\end{bmatrix}.}}}$For rotated MC-CDM, the combined frequency-domain channel responsematrix can be as shown in Equation 5. $\begin{matrix}{{\hat{H}\left( \theta_{1} \right)} = {{\overset{\sim}{H}{R_{2}\left( \theta_{1} \right)}} = {{\begin{bmatrix}{\overset{\sim}{h}}_{1} & \quad \\\quad & {\overset{\sim}{h}}_{2}\end{bmatrix}\begin{bmatrix}{\cos\left( \theta_{1} \right)} & {\sin\left( \theta_{1} \right)} \\{- {\sin\left( \theta_{1} \right)}} & {\cos\left( \theta_{1} \right)}\end{bmatrix}} = \begin{bmatrix}{{\overset{\sim}{h}}_{1}{\cos\left( \theta_{1} \right)}} & {{\overset{\sim}{h}}_{1}{\sin\left( \theta_{1} \right)}} \\{{- {\overset{\sim}{h}}_{2}}{\sin\left( \theta_{1} \right)}} & {{\overset{\sim}{h}}_{2}{\cos\left( \theta_{1} \right)}}\end{bmatrix}}}} & \left\lbrack {{Equation}\quad 5} \right\rbrack\end{matrix}$

The effect of the transform can be illustrated in a correlation matrixof Equation 6. $\begin{matrix}\begin{matrix}{C = {{\hat{H}}^{- H}H}} \\{= {\begin{bmatrix}{{\overset{\sim}{h}}_{1}^{*}\alpha^{*}} & {{- {\overset{\sim}{h}}_{2}^{*}}\beta} \\{{\overset{\sim}{h}}_{1}^{*}\beta^{*}} & {{\overset{\sim}{h}}_{2}^{*}\alpha^{*}}\end{bmatrix}\begin{bmatrix}{{\overset{\sim}{h}}_{1}\alpha} & {{\overset{\sim}{h}}_{1}\beta} \\{{- {\overset{\sim}{h}}_{2}}\beta^{*}} & {{\overset{\sim}{h}}_{2}\alpha^{*}}\end{bmatrix}}} \\{= \begin{bmatrix}{{{{\overset{\sim}{h}}_{1}}^{2}{\alpha }^{2}} + {{{\overset{\sim}{h}}_{2}}{\beta }^{2}}} & {\left( {{{\overset{\sim}{h}}_{1}}^{2} - {{\overset{\sim}{h}}_{2}}^{2}} \right)\alpha^{*}\beta} \\{\left( {{- {{\overset{\sim}{h}}_{1}}^{2}} - {{\overset{\sim}{h}}_{2}}^{2}} \right){\alpha\beta}^{*}} & {{{{\overset{\sim}{h}}_{2}}^{2}{\alpha }^{2}} + {{{\overset{\sim}{h}}_{1}}^{2}{\beta }^{2}}}\end{bmatrix}} \\{= {D + I}} \\{= {\begin{bmatrix}{{{{\overset{\sim}{h}}_{1}}^{2}{\alpha }^{2}} + {{{\overset{\sim}{h}}_{2}}^{2}{\beta }^{2}}} & 0 \\0 & {{{{\overset{\sim}{h}}_{2}}^{2}{\alpha }^{2}} + {{{\overset{\sim}{h}}_{1}}^{2}{\beta }^{2}}}\end{bmatrix} +}} \\{\begin{bmatrix}0 & {\left( {{{\overset{\sim}{h}}_{1}}^{2} - {{\overset{\sim}{h}}_{2}}^{2}} \right)\alpha^{*}\beta} \\{\left( {{- {{\overset{\sim}{h}}_{1}}^{2}} - {{\overset{\sim}{h}}_{2}}^{2}} \right){\alpha\beta}^{*}} & 0\end{bmatrix}}\end{matrix} & \left\lbrack {{Equation}\quad 6} \right\rbrack\end{matrix}$

Referring to Equation 3 the diversity can be denoted by${D = \begin{bmatrix}{{{{\overset{\sim}{h}}_{1}}^{2}{\alpha }^{2}} + {{{\overset{\sim}{h}}_{2}}^{2}{\beta }^{2}}} & 0 \\0 & {{{{\overset{\sim}{h}}_{2}}^{2}{\alpha }^{2}} + {{{\overset{\sim}{h}}_{1}}^{2}{\beta }^{2}}}\end{bmatrix}},$and the interference matrix can be denoted${{by}\quad I} = {\begin{bmatrix}0 & {\left( {{{\overset{\sim}{h}}_{1}}^{2} - {{\overset{\sim}{h}}_{2}}^{2}} \right)\alpha^{*}\beta} \\{\left( {{- {{\overset{\sim}{h}}_{1}}^{2}} + {{\overset{\sim}{h}}_{2}}^{2}} \right){\alpha\beta}^{*}} & 0\end{bmatrix}.}$Here, the interference matrix can be ISI or multiple access interference(MAI).

A total diversity of the generalized MC-CDM can be represented as shownin Equation 7. $\begin{matrix}\begin{matrix}{D = {{Tr}\left\{ D \right\}}} \\{= {{Tr}\left\{ \begin{bmatrix}{{{{\overset{\sim}{h}}_{1}}^{2}{\alpha }^{2}} + {{{\overset{\sim}{h}}_{2}}^{2}{\beta }^{2}}} & 0 \\0 & {{{{\overset{\sim}{h}}_{2}}^{2}{\alpha }^{2}} + {{{\overset{\sim}{h}}_{1}}^{2}{\beta }^{2}}}\end{bmatrix} \right\}}} \\{= {{{\overset{\sim}{h}}_{1}}^{2} + {{\overset{\sim}{h}}_{2}}^{2}}}\end{matrix} & \left\lbrack {{Equation}\quad 7} \right\rbrack\end{matrix}$

Referring to Equation 4, the total diversity of the generalized MC-CDMis independent on the precoding matrix U. However, for each symbol oruser, the diversity gain may be different to each.

Further, the interference of the generalized MC-CDM can be representedas shown in Equation 8. $\begin{matrix}\begin{matrix}{I = {{Tr}_{2}\left\{ I \right\}}} \\{= {{Tr}_{2}\left\{ \begin{bmatrix}0 & {\left( {{{\overset{\sim}{h}}_{1}}^{2} - {{\overset{\sim}{h}}_{2}}^{2}} \right)\alpha^{*}\beta} \\{\left( {{- {{\overset{\sim}{h}}_{1}}^{2}} + {{\overset{\sim}{h}}_{2}}^{2}} \right){\alpha\beta}^{*}} & 0\end{bmatrix} \right\}}} \\{= {{2{{{{\overset{\sim}{h}}_{1}}^{2} - {{\overset{\sim}{h}}_{2}}^{2}}}{{\alpha\beta}^{*}}} \leq {{{{\overset{\sim}{h}}_{1}}^{2} - {{\overset{\sim}{h}}_{2}}^{2}}}}}\end{matrix} & \left\lbrack {{Equation}\quad 8} \right\rbrack\end{matrix}$

Here, if |{tilde over (h)}₁|²≠|{tilde over (h)}₂|² and |αβ*|≠0, there issome self-interference or multi-user. interference. In other words, dueto frequency-selectivity in OFDM-liked orthogonal modulation, there ispossible interference if some preceding or spreading is applied.Furthermore, it can be shown that this interference can be maximizedwhen the rotation ${{angel}\quad{is}\quad\theta} = {\frac{\pi}{4}.}$

In designing a MC-CDM transceiver, inter alia, an inter-symbol ormultiple access signal-to-interference ratio (SIR) can be defined asfollows. $\begin{matrix}{{SIR}_{1} = {\frac{{{{\overset{\sim}{h}}_{1}}^{2}{\alpha }^{2}} + {{{\overset{\sim}{h}}_{2}}^{2}{\beta }^{2}}}{{{{{\overset{\sim}{h}}_{1}}^{2} - {{\overset{\sim}{h}}_{2}}^{2}}}{{\alpha\beta}^{*}}} = \frac{{\alpha }^{2} + {\gamma{\beta }^{2}}}{{{1 - \gamma}}{{\alpha\beta}^{*}}}}} & \left\lbrack {{Equation}\quad 9} \right\rbrack\end{matrix}$

Referring to Equation 9,$\gamma = \frac{{{\overset{\sim}{h}}_{2}}^{2}}{{{\overset{\sim}{h}}_{1}}^{2}}$denotes the channel fading difference. The SIR can be defined based onchannel fading and rotation.

Rotation can also be performed based on receiver profile. This can bedone through upper layer signaling. More specifically, at least twoparameters can be configured, namely, spreading gain and rotation angle.

In operation, a receiver can send feedback information containing itsoptimum rotation angle and/or rotation index. The rotation angle and/orrotation index can be mapped to the proper rotation angle by atransmitter based on a table (or index). This table or index is known byboth the transmitter and the receiver. This can be done any time when itis the best time for the transmitter and/or receiver.

For example, if the receiver (or access terminal) is registered with thenetwork, it usually sends its profile to the network. This profileincludes, inter alia, the rotation angle and/or index.

Before the transmitter decides to send signals to the receiver, it mayask the receiver as to the best rotation angle. In response, thereceiver can send the best rotation angle to the transmitter.Thereafter, the transmitter can send the signals based on the feedbackinformation and its own decision.

During transmission of the signals, the transmitter can periodicallyrequest from the receiver to send its updated rotation angle.Alternatively, the transmitter can request an update of the rotationangle from the receiver after the transmitter is finished transmitting.

At any time, the receiver can send the update (or updated rotationangle) to the transmitter. The transmission of the update (or feedbackinformation) can be executed through an access channel, traffic channel,control channel, or other possible channels.

With respect to channel coding, coding can help minimize demodulationerrors and therefore achieve the throughput in addition to signal designfor higher spectral efficiency. In reality, most capacity-achievingcodes are designed to balance the implementation complexity andachievable performance.

Gray code is one of an example of channel coding which is also known asreflective binary code. Gray code or the reflective binary code is abinary numeral system where two successive values differ in only onedigit. FIG. 7 is an exemplary diagram illustrating an example of Graycoding.

Gray code for bits-to-symbol mapping, also called Gray mapping, can beimplemented with other channel coding scheme. Gray mapping is generallyaccepted as the optimal mapping rule for minimizing bit error rate(BER). Gray mapping for regular QPSK/QPSK hierarchical modulation (or 16QAM modulation) is shown in FIG. 8 where the codewords with minimumEuclid distance have minimum Hamming distance as well.

In the figures to follow, the Gray mapping rule is described. Morespecifically, each enhancement layer bits-to-symbol and base layerbits-to-symbol satisfy the Gray mapping requirement where the closesttwo symbols only have difference of one or the least bit(s).Furthermore, the overall bits-to-symbol mapping rule satisfies the Graymapping rule.

FIG. 8 is an exemplary diagram illustrating mapping for regularQPSK/QPSK hierarchical modulation or 16 QAM modulation. Referring toFIG. 8, the enhancement layer bits and the base layer bits can bearbitrarily combined so that every time when the base layer bits aredetected, the enhancement layer bits-to-symbol mapping table/rule can bedecided, for example. In addition, both the base layer and theenhancement layer are QPSK. Furthermore, every point (or symbol) isrepresented and/or mapped by b₀b₁b₂b₃.

More specifically, the circle in the center of the diagram and the linesconnecting two (2) points (or symbols) (e.g., point 0011 and point 0001or point 0110 and point 1110) represent connection with only one bitdifference between neighbors. Here, the connected points are fromdifferent layers. In other words, every connected points (or symbol) aredifferent base layer bits and enhancement layer bits.

Furthermore, every point can be represented by four (4) bits (e.g.,b₀b₁b₂b₃) in which the first bit (b₀) and the third bit (b₂) representthe base layer bits, and the second bit (b₁) and the fourth bit (b₃)represent the enhancement bits. That is, two (2) bits from the baselayer and the two (2) bits from the enhancement layer are interleavedtogether to represent every resulted point. By interleaving the bitsinstead of simple concatenation of the bits from two layers, additionaldiversity gain can be potentially attained.

FIG. 9 is an exemplary diagram illustrating bits-to-symbol mapping for16 QAM/QPSK. This figure refers to bits-to-symbol mapping. This mappingcan be used by both the transmitter and the receiver.

If a transmitter desires to send bits b₀b₁b₂b₃b₄b₅, the transmitterneeds to look for a mapped symbol to send. Hence, if a receiver desiresto demodulate the received symbol, the receiver can use this figure tofind/locate the demodulated bits.

Furthermore, FIG. 9 represents 16 QAM/QPSK hierarchical modulation. Inother words, the base layer is modulated by 16 QAM, and the enhancementlayer is modulated by QPSK. Moreover, 16 QAM/QPSK can be referred to asa special hierarchical modulation. In other words, the base layer signaland the enhancement signal have different initial phase. For example,the base layer signal phase is 0 while the enhancement layer signalphase is theta (θ).

Every symbol in FIG. 9 is represented by bits sequence, s₅s₄s₃s₁S₁s₀, inwhich bits S₃ and s₀ are bits from the enhancement layer while the otherbits (e.g., s₅, s₄, s₂, and s₁) belong to the base layer.

FIG. 10 is another exemplary diagram illustrating bits-to-symbol mappingfor 16 QAM/QPSK. The difference between FIG. 10 and previous FIG. 9 isthat every symbol in FIG. 10 is represented by bits sequence,s₅s₄s₃s₂s₁s₀ in which bits s₅ and s₂ are bits from the enhancement layerwhile the other bits (e.g., s₄, s₃, s₁, and s₀) are from the base layer.

FIG. 11 is another exemplary diagram illustrating bits-to-symbol mappingfor 16 QAM/QPSK. The difference between FIG. 11 and previous FIGS. 9and/or 10 is that every symbol in FIG. 11 is represented by bitssequence, s₅s₄s₃s₂s₁s₀, in which bits s₅ and s₄ are bits from theenhancement layer while the other bits (e.g., s₃, s₂, s₁, and s₀) arefrom the base layer.

FIG. 12 is another exemplary diagram illustrating bits-to-symbol mappingfor 16 QAM/QPSK. The difference between FIG. 12 and previous FIGS. 9,10, and/or 11 bits s₅ and s₂ are bits from the enhancement layer whilethe other bits (e.g., s₄, s₃, s₁, and s₀) are from the base layer. Asbefore, every symbol in FIG. 12 is represented by bits sequence,s₅s₄s₃s₂s₁s₀.

Further to bits sequence combinations as discussed above, the followinghierarchical layer and enhancement layer combination possibilitiesinclude (1) s₅s₄s₃s₂s₁s₀=b₃b₂b₁e₁b₀e₀, (2) s₅s₄s₃s₂s₁s₀=b₃e₁b₂b₁b₀e₀,(3) s₅s₄s₃s₂s₁s₀=b₃b₂b₁b₀e₀e₁, (4) s₅s₄s₃s₂s₁s₀=e₀e₁b₃b₂b₁b₀, (5)s₅s₄s₃s₂s₁S₀=e₀b₃b₂e₁b₁b₀, (6) s₅s₄s₃s₂s₁s₀=b₃b₂e₀b₁b₀e_(l), (7)s₃s₂s₀=e₁b_(l)e₀b₀, (8) s₃s₂s₁s₀=e₀b₁e_(l)b₀, (9) s₃s₂s₁s₀=e₁e₀b₁b₀,(10) s₃s₂s₁s₀=e₀e₁b₁b₀, and (11) s₃s₂s₁s₀=b₁b₀e₀e₁.

In addition to the combinations discussions of above, there are manyother possible combinations. However, they all follow the same rulewhich is the Gray rule or the Gray mapping rule. As discussed, eachenhancement layer bits-to-symbol mapping and base layer bits-to-symbolmapping satisfy the Gray mapping rule requirement which is that theclosest two symbols only have difference of one bit or less. Moreover,the overall bits-to-symbol mapping rule satisfies the Gray mapping ruleas well.

Further, the enhancement layer bits and the base layer bits can bearbitrarily combined so that every time the base layer bits aredetected, the enhancement layer bits-to-symbol mapping table/rule can bedecided, In addition, it is possible, for example, for s₃s₂s₁s₀=e₁e₀b₁b₀QPSK/QPSK, the Gray mapping rule for enhancement layer s₃s₂11=e₁e₀11 tobe not the exactly the same as s₃s₂10=e_(l)e₀10. Moreover, for example,it is possible s₃s₂11=e₁e₀11 is a rotated version ass₃s₂10=e₁e₀10=e₁e₀11=1111's position is the position of s₃s₂11=1010 ors₃s₂11=0110.

FIG. 13 is an exemplary diagram illustrating bits-to-symbol mapping forQPSK/QPSK. Referring to FIG. 13, the bits-to-symbol mapping can be usedby both the transmitter and the receiver. If a transmitter desires tosend bits b₀b₁b₂b₃, the transmitter needs to look for a mapped symbol tosend. Hence, if a receiver desires to demodulate the received symbol,the receiver can use this figure to find/locate the demodulated bits.

Furthermore, FIG. 13 represents QPSK/QPSK hierarchical modulation. Inother words, the base layer is modulated by QPSK, and the enhancementlayer is also modulated by QPSK. Moreover, QPSK/QPSK can be referred toas a special hierarchical modulation. That is, the base layer signal andthe enhancement signal have different initial phase. For example, thebase layer signal phase is 0 while the enhancement layer signal phase istheta (θ).

Every symbol in FIG. 13 is represented by bits sequence, s₃s₂s₁s₀, inwhich bits s₃ and s₁ are bits from the enhancement layer while the otherbits (e.g., S₂ and s₀) belong to the base layer.

Further, in the QPSK/QPSK example, the enhancement layer bits-to-symbolmapping rules may be different from the base layer symbol-to-symbol.FIG. 14 is an exemplary diagram illustrating an enhancement layerbits-to-symbol for base layer 0x0. In other words, FIG. 14 illustratesan example of how the base layer bits are mapped.

For example, the symbols indicated in the upper right quadrant denotethe base layer symbols of ‘00’. This means that as long as the baselayer bits are ‘00’, whatever the enhancement layer is, thecorresponding layer modulated symbol is one of the four (4) symbols ofthis quadrant.

FIG. 15 is an exemplary diagram illustrating an enhancement layerbits-to-symbol for base layer 0x1. Similarly, this diagram illustratesanother example of how the base layer bits are mapped. For example, thesymbols of in the upper left quadrant denote the base layer symbols of‘01’. This means that as long as the base layer bits are ‘01’, whateverthe enhancement layer bits are, the corresponding layer modulatedsymbols is one of the symbols of the upper left quadrant.

As discussed above with respect to FIGS. 1-3, the inputted data or datastream can be channel coded using the Gray mapping rule, for example,followed by other processes including modulation. The modulationdiscussed here refers to layered (or superposition) modulation. Thelayered modulation is a type of modulation in which each modulationsymbol has bits corresponding to both a base layer and an enhancementlayer. In the discussions to follow, the layered modulation will bedescribed in the context of broadcast and multicast services (BCMCS).

In general, layered modulation can be a superposition of any twomodulation schemes. In BCMCS, a QPSK enhancement layer is superposed ona base QPSK or 16-QAM layer to obtain the resultant signalconstellation. The energy ratio r is the power ratio between the baselayer and the enhancement. Furthermore, the enhancement layer is rotatedby the angle θ in counter-clockwise direction.

FIG. 16 is an exemplary diagram showing the signal constellation of thelayered modulator with respect to QPSK/QPSK hierarchical modulation.Referring to QPSK/QPSK hierarchical modulation, which means QPSK baselayer and QPSK enhancement layer, each modulation symbol contains four(4) bits, namely, s₃, s₂, s₁, s₀. Here, there are two (2) mostsignificant bits (MSBs) which are s₃ and s₂, and two (2) leastsignificant bits (LSBs) which are s₁ and s₀. The two (2) MSBs are fromthe base layer and the two LSBs come from the enhancement layer.

Given energy ratio r between the base layer and enhancement layer,$\alpha = {{\sqrt{\frac{r}{2\left( {1 + r} \right)}}\quad{and}\quad\beta} = \sqrt{\frac{1}{2\left( {1 + r} \right)}}}$can be defined such that 2(α²+β²)=1. Here, a denotes the amplitude ofthe base layer, and β denotes the amplitude of enhancement layer.Moreover, 2(α²+β²)=1 is a constraint which is also referred to as powerconstraint and more accurately referred to as normalization.

Table 1 illustrates a layered modulation table with QPSK base layer andQPSK enhancement layer. TABLE 1 Modulator Input Bits Modulation Symbolss₃ s₂ s₁ s₀ m_(I)(k) m_(Q)(k) 0 0 0 0 α + {square root over (2)} cos(θ +π/4)β α + {square root over (2)} sin(θ + π/4)β 0 0 0 1 α + {square rootover (2)} cos(θ + 3π/4)β α + {square root over (2)} sin(θ + 3π/4)β 0 1 01 α + {square root over (2)} cos(θ + 7π/4)β α + {square root over (2)}sin(θ + 7π/4)β 0 1 0 0 α + {square root over (2)} cos(θ + 5π/4)β α +{square root over (2)} sin(θ + 5π/4)β 0 0 1 1 −α + {square root over(2)} cos(θ + π/4)β α + {square root over (2)} sin(θ + π/4)β 0 0 1 0 −α +{square root over (2)} cos(θ + 3π/4)β α + {square root over (2)} sin(θ +3π/4)β 0 1 1 0 −α + {square root over (2)} cos(θ + 7π/4)β α + {squareroot over (2)} sin(θ + 7π/4)β 0 1 1 1 −α + {square root over (2)}cos(θ + 5π/4)β α + {square root over (2)} sin(θ + 5π/4)β 1 1 0 0 α +{square root over (2)} cos(θ + π/4)β −α + {square root over (2)} sin(θ +π/4)β 1 1 0 1 α + {square root over (2)} cos(θ + 3π/4)β −α + {squareroot over (2)} sin(θ + 3π/4)β 1 0 0 1 α + {square root over (2)} cos(θ +7π/4)β −α + {square root over (2)} sin(θ + 7π/4)β 1 0 0 0 α + {squareroot over (2)} cos(θ + 5π/4)β −α + {square root over (2)} sin(θ + 5π/4)β1 1 1 1 −α + {square root over (2)} cos(θ + π/4)β −α + {square root over(2)} sin(θ + π/4)β 1 1 1 0 −α + {square root over (2)} cos(θ + 3π/4)β−α + {square root over (2)} sin(θ + 3π/4)β 1 0 1 0 −α + {square rootover (2)} cos(θ + 7π/4)β −α + {square root over (2)} sin(θ + 7π/4)β 1 01 1 −α + {square root over (2)} cos(θ + 5π/4)β −α + {square root over(2)} sin(θ + 5π/4)β

Referring to Table 1, each column defines the symbol position for eachfour (4) bits, s₃, s₂, s₁, s₀. Here, the position of each symbol isgiven in a two-dimensional signal space (m₁, m_(Q). This means that eachsymbol can be represented by S(t)=└M₁ cos(2πf₀t+φ₀)+*sin(2πf₀t+φ₀)┘φ(1).Simply put, the complex modulation symbol S=r (m₁, m_(Q)) for each [s₃,s₂, s₁, s₀] is specified in S(t)=└M₁cos(2πf₀+φ₀)+M_(Q)*sin(2πf₀t+φ₀)┘φ(t).

Here, cos(2πf₀t+φ₀) and sin(2πf₀t+φ₀) denote the carrier signal withinitial phase φ₀ and carrier frequency f₀. Moreover, φ(t) denotes thepulse-shaping, the shape of a transmit symbol.

In the above definition of S(t), except the m₁ and m_(Q) value, otherparameters can usually either be shared between the transmitter and thereceiver or be detected by the receiver itself. For correctlydemodulating S(t), it is necessary to define and share the possiblevalue information of m₁ and m_(Q).

The possible value of m₁(k) and m_(Q)(k), which denote the m₁ and m_(Q)value for the k^(th) symbol, are given in Table 1. It shows forrepresenting each group inputs bits s₃, s₂, s₁, s₀ the symbol shall bemodulated by corresponding parameters shown in the table.

The discussion with respect to the complex modulation symbol can beapplied in a similar or same manner to the following discussions ofvarious layered modulations. That is, the above discussion of thecomplex modulation symbol can be applied to the tables to follow.

FIG. 17 is an exemplary diagram illustrating the signal constellation ofthe layered modulator with respect to 16 QAM/QPSK hierarchicalmodulation. Referring to 16 QAM/QPSK hierarchical modulation, whichmeans 16 QAM base layer and QPSK enhancement layer, each modulationsymbol contains six 6 bits—s₅, s₄, s₃, s₂, s₁, s₀. The four (4) MSBs,s₅, s₄, s₃ and s₂, come from the base layer, and the two (2) LSBs, s₁and s₀, come from the enhancement layer.

Given energy ratio r between the base layer and enhancement layer,$\alpha = {{\sqrt{\frac{r}{2\left( {1 + r} \right)}}\quad{and}\quad\beta} = \sqrt{\frac{1}{2\left( {1 + r} \right)}}}$can be defined such that 2(α²+β²)=1. Here, α denotes the amplitude ofthe base layer, and β denotes the amplitude of enhancement layer.Moreover, 2(α²+β²)=1 is a constraint which is also referred to as powerconstraint and more accurately referred to as normalization.

Table 2 illustrates a layered modulation table with 16 QAM base layerand QPSK enhancement layer. TABLE 2 Modulator Input Bits ModulationSymbols s₅ s₄ s₃ s₂ s₁ s₀ m_(I)(k) m_(Q)(k) 0 0 0 0 0 0 3α + {squareroot over (2)} cos(θ + π/4)β 3α + {square root over (2)} sin(θ + π/4)β 00 0 0 0 1 3α + {square root over (2)} cos(θ + 3π/4)β 3α + {square rootover (2)} sin(θ + 3π/4)β 0 0 1 0 0 1 3α + {square root over (2)} cos(θ +7π/4)β 3α + {square root over (2)} sin(θ + 7π/4)β 0 0 1 0 0 0 3α +{square root over (2)} cos(θ + 5π/4)β 3α + {square root over (2)}sin(θ + 5π/4)β 0 0 0 0 1 1 α + {square root over (2)} cos(θ + π/4)β 3α +{square root over (2)} sin(θ + π/4)β 0 0 0 0 1 0 α + {square root over(2)} cos(θ + 3π/4)β 3α + {square root over (2)} sin(θ + 3π/4)β 0 0 1 0 10 α + {square root over (2)} cos(θ + 7π/4)β 3α + {square root over (2)}sin(θ + 7π/4)β 0 0 1 0 1 1 α + {square root over (2)} cos(θ + 5π/4)β3α + {square root over (2)} sin(θ + 5π/4)β 0 0 0 1 0 1 −3α + {squareroot over (2)} cos(θ + π/4)β 3α + {square root over (2)} sin(θ + π/4)β 00 0 1 0 0 −3α + {square root over (2)} cos(θ + 3π/4)β 3α + {square rootover (2)} sin(θ + 3π/4)β 0 0 1 1 0 0 −3α + {square root over (2)}cos(θ + 7π/4)β 3α + {square root over (2)} sin(θ + 7π/4)β 0 0 1 1 0 1−3α + {square root over (2)} cos(θ + 5π/4)β 3α + {square root over (2)}sin(θ + 5π/4)β 0 0 0 1 1 0 −α + {square root over (2)} cos(θ + π/4)β3α + {square root over (2)} sin(θ + π/4)β 0 0 0 1 1 1 −α + {square rootover (2)} cos(θ + 3π/4)β 3α + {square root over (2)} sin(θ + 3π/4)β 0 01 1 1 1 −α + {square root over (2)} cos(θ + 7π/4)β 3α + {square rootover (2)} sin(θ + 7π/4)β 0 0 1 1 1 0 −α + {square root over (2)} cos(θ +5π/4)β 3α + {square root over (2)} sin(θ + 5π/4)β 0 1 1 0 0 0 3α +{square root over (2)} cos(θ + π/4)β α + {square root over (2)} sin(θ +π/4)β 0 1 1 0 0 1 3α + {square root over (2)} cos(θ + 3π/4)β α + {squareroot over (2)} sin(θ + 3π/4)β 0 1 0 0 0 1 3α + {square root over (2)}cos(θ + 7π/4)β α + {square root over (2)} sin(θ + 7π/4)β 0 1 0 0 0 03α + {square root over (2)} cos(θ + 5π/4)β α + {square root over (2)}sin(θ + 5π/4)β 0 1 1 0 1 1 α + {square root over (2)} cos(θ + π/4)β α +{square root over (2)} sin(θ + π/4)β 0 1 1 0 1 0 α + {square root over(2)} cos(θ + 3π/4)β α + {square root over (2)} sin(θ + 3π/4)β 0 1 0 0 10 α + {square root over (2)} cos(θ + 7π/4)β α + {square root over (2)}sin(θ + 7π/4)β 0 1 0 0 1 1 α + {square root over (2)} cos(θ + 5π/4)β α +{square root over (2)} sin(θ + 5π/4)β 0 1 1 1 0 1 −3α + {square rootover (2)} cos(θ + π/4)β α + {square root over (2)} sin(θ + π/4)β 0 1 1 10 0 −3α + {square root over (2)} cos(θ + 3π/4)β α + {square root over(2)} sin(θ + 3π/4)β 0 1 1 1 0 1 −3α + {square root over (2)} cos(θ +7π/4)β α + {square root over (2)} sin(θ + 7π/4)β 0 1 0 1 0 1 −3α +{square root over (2)} cos(θ + 5π/4)β α + {square root over (2)} sin(θ +5π/4)β 0 1 1 1 1 0 −α + {square root over (2)} cos(θ + π/4)β α + {squareroot over (2)} sin(θ + π/4)β 0 1 1 1 1 1 −α + {square root over (2)}cos(θ + 3π/4)β α + {square root over (2)} sin(θ + 3π/4)β 0 1 0 1 1 1−α + {square root over (2)} cos(θ + 7π/4)β α + {square root over (2)}sin(θ + 7π/4)β 0 1 0 1 1 0 −α + {square root over (2)} cos(θ + 5π/4)βα + {square root over (2)} sin(θ + 5π/4)β 1 0 1 0 0 0 3α + {square rootover (2)} cos(θ + π/4)β −3α + {square root over (2)} sin(θ + π/4)β 1 0 10 0 1 3α + {square root over (2)} cos(θ + 3π/4)β −3α + {square root over(2)} sin(θ + 3π/4)β 1 0 0 0 0 1 3α + {square root over (2)} cos(θ +7π/4)β −3α + {square root over (2)} sin(θ + 7π/4)β 1 0 0 0 0 0 3α +{square root over (2)} cos(θ + 5π/4)β −3α + {square root over (2)}sin(θ + 5π/4)β 1 0 1 0 1 1 α + {square root over (2)} cos(θ + π/4)β−3α + {square root over (2)} sin(θ + π/4)β 1 0 1 0 1 0 α + {square rootover (2)} cos(θ + 3π/4)β −3α + {square root over (2)} sin(θ + 3π/4)β 1 00 0 1 0 α + {square root over (2)} cos(θ + 7π/4)β −3α + {square rootover (2)} sin(θ + 7π/4)β 1 0 0 0 1 1 α + {square root over (2)} cos(θ +5π/4)β −3α + {square root over (2)} sin(θ + 5π/4)β 1 0 1 1 0 1 −3α +{square root over (2)} cos(θ + π/4)β −3α + {square root over (2)}sin(θ + π/4)β 1 0 1 1 0 0 −3α + {square root over (2)} cos(θ + 3π/4)β−3α + {square root over (2)} sin(θ + 3π/4)β 1 0 0 1 0 0 −3α + {squareroot over (2)} cos(θ + 7π/4)β −3α + {square root over (2)} sin(θ +7π/4)β 1 0 0 1 0 1 −3α + {square root over (2)} cos(θ + 5π/4)β −3α +{square root over (2)} sin(θ + 5π/4)β 1 0 1 1 1 0 −α + {square root over(2)} cos(θ + π/4)β −3α + {square root over (2)} sin(θ + π/4)β 1 0 1 1 11 −α + {square root over (2)} cos(θ + 3π/4)β −3α + {square root over(2)} sin(θ + 3π/4)β 1 0 0 1 1 1 −α + {square root over (2)} cos(θ +7π/4)β −3α + {square root over (2)} sin(θ + 7π/4)β 1 0 0 1 1 0 −α +{square root over (2)} cos(θ + 5π/4)β −3α + {square root over (2)}sin(θ + 5π/4)β 1 1 0 1 0 1 3α + {square root over (2)} cos(θ + π/4)β−α + {square root over (2)} sin(θ + π/4)β 1 1 0 1 0 0 3α + {square rootover (2)} cos(θ + 3π/4)β −α + {square root over (2)} sin(θ + 3π/4)β 1 11 1 0 0 3α + {square root over (2)} cos(θ + 7π/4)β −α + {square rootover (2)} sin(θ + 7π/4)β 1 1 1 1 0 1 3α + {square root over (2)} cos(θ +5π/4)β −α + {square root over (2)} sin(θ + 5π/4)β 1 1 0 1 1 0 α +{square root over (2)} cos(θ + π/4)β −α + {square root over (2)} sin(θ +π/4)β 1 1 0 1 1 1 α + {square root over (2)} cos(θ + 3π/4)β −α + {squareroot over (2)} sin(θ + 3π/4)β 1 1 1 1 1 1 α + {square root over (2)}cos(θ + 7π/4)β −α + {square root over (2)} sin(θ + 7π/4)β 1 1 1 1 1 0α + {square root over (2)} cos(θ + 5π/4)β −α + {square root over (2)}sin(θ + 5π/4)β 1 1 0 1 0 1 −3α + {square root over (2)} cos(θ + π/4)β−α + {square root over (2)} sin(θ + π/4)β 1 1 0 1 0 0 −3α + {square rootover (2)} cos(θ + 3π/4)β −α + {square root over (2)} sin(θ + 3π/4)β 1 11 1 0 0 −3α + {square root over (2)} cos(θ + 7π/4)β −α + {square rootover (2)} sin(θ + 7π/4)β 1 1 1 1 0 1 −α + {square root over (2)} cos(θ +5π/4)β −α + {square root over (2)} sin(θ + 5π/4)β 1 1 0 1 1 0 −α +{square root over (2)} cos(θ + π/4)β −α + {square root over (2)} sin(θ +π/4)β 1 1 0 1 1 1 −α + {square root over (2)} cos(θ + 3π/4)β −α +{square root over (2)} sin(θ + 3π/4)β 1 1 1 1 1 1 −α + {square root over(2)} cos(θ + 7π/4)β −α + {square root over (2)} sin(θ + 7π/4)β 1 1 1 1 10 −α + {square root over (2)} cos(θ + 5π/4)β −α + {square root over (2)}sin(θ + 5π/4)β

Referring to Table 2, each column defines the symbol position for eachsix (6) bits, s₅, s₄, s₃, s₂, s₁, s₀. Here, the position of each symbolis given in a two-dimensional signal space (m₁, m_(Q)). This means thateach symbol can be represented by S(t) └M₁cos(2πf₀t+φ₀)+M_(Q)*sin(2πf₀t+φ₀)┘φ(t). Simply put, the complexmodulation symbol S=(m₁, m_(Q)) for each [s₅, s₄, s₃, s₂, s₁, s₀] isspecified in S(t)=└M₁ cos(2πf₀t+φ₀))+M_(Q)*sin(2πf₀t+φ₀)┘φ(t).

Here, w₀ denotes carrier frequency, π₀ denotes an initial phase of thecarrier, and φ(t) denotes the symbol shaping or pulse shaping wave.Here, cos(2πf₀t+φ₀) and sin(2πf₀t+φ₀) denote the carrier signal withinitial phase φ₀ and carrier frequency f₀ Moreover, φ(t) denotes thepulse-shaping, the shape of a transmit symbol.

In the above definition of S(t), except the m₁ and m_(Q) value, otherparameters can usually either be shared between the transmitter and thereceiver or be detected by the receiver itself. For correctlydemodulating S(i), it is necessary to define and share the possiblevalue information of m₁ and m_(Q).

The possible value of m₁(k) and m_(Q)(k), which denote the m₁ and m_(Q)value for the k^(th) symbol, are given in Table 1. It shows forrepresenting each group inputs bits s₅, s₄, s₃, s₂, s₁, s₀ the symbolshall be modulated by corresponding parameters shown in the table.

Further, another application example for BCMCS for hierarchicalmodulation is discussed below. In general, layered modulation can be asuperposition of any two modulation schemes. In BCMCS, a QPSKenhancement layer is superposed on a base QPSK or 16-QAM layer to obtainthe resultant signal constellation. The energy ratio r is the powerratio between the base layer and the enhancement. Furthermore, theenhancement layer is rotated by the angle θ in counter-clockwisedirection.

FIG. 18 is an exemplary diagram showing the signal constellation for thelayered modulator with QPSK/QPSK hierarchical modulation. Referring toQPSK/QPSK hierarchical modulation, which means QPSK base layer and QPSKenhancement layer, each modulation symbol contains four (4) bits,namely, s₃, s₂, s₁, s₀. Here, there are two (2) MSBs which are s₃ ands₂, and two (2) LSBs which are s₁ and s₀. The two (2) MSBs are from thebase layer and the two LSBs come from the enhancement layer.

Given energy ratio r between the base layer and enhancement layer,$\alpha = {{\sqrt{\frac{r}{2\left( {1 + r} \right)}}\quad{and}\quad\beta} = \sqrt{\frac{1}{2\left( {1 + r} \right)}}}$can be defined such that 2(α²+β²)=1. Here, a denotes the amplitude ofthe base layer, and p denotes the amplitude of enhancement layer.Moreover, 2(α²+β²)=1 is a constraint which is also referred to as powerconstraint and more accurately referred to as normalization.

Table 3 illustrates a layered modulation table with QPSK base layer andQPSK enhancement layer. TABLE 3 Modulator Input Bits Modulation Symbolss₃ s₂ s₁ s₀ m_(I)(k) m_(Q)(k) 0 0 0 0 α + {square root over (2)} cos(θ +π/4)β α + {square root over (2)} sin(θ + π/4)β 0 0 1 0 α + {square rootover (2)} cos(θ + 3π/4)β α + {square root over (2)} sin(θ + 3π/4)β 1 0 00 α + {square root over (2)} cos(θ + 7π/4)β α + {square root over (2)}sin(θ + 7π/4)β 1 0 1 0 α + {square root over (2)} cos(θ + 5π/4)β α +{square root over (2)} sin(θ + 5π/4)β 0 0 1 1 −α + {square root over(2)} cos(θ + π/4)β α + {square root over (2)} sin(θ + π/4)β 0 0 0 1 −α +{square root over (2)} cos(θ + 3π/4)β α + {square root over (2)} sin(θ +3π/4)β 1 0 1 1 −α + {square root over (2)} cos(θ + 7π/4)β α + {squareroot over (2)} sin(θ + 7π/4)β 1 0 0 1 −α + {square root over (2)}cos(θ + 5π/4)β α + {square root over (2)} sin(θ + 5π/4)β 1 1 0 0 α +{square root over (2)} cos(θ + π/4)β −α + {square root over (2)} sin(θ +π/4)β 1 1 1 0 α + {square root over (2)} cos(θ + 3π/4)β −α + {squareroot over (2)} sin(θ + 3π/4)β 0 1 0 0 α + {square root over (2)} cos(θ +7π/4)β −α + {square root over (2)} sin(θ + 7π/4)β 0 1 1 0 α + {squareroot over (2)} cos(θ + 5π/4)β −α + {square root over (2)} sin(θ + 5π/4)β1 1 1 1 −α + {square root over (2)} cos(θ + π/4)β −α + {square root over(2)} sin(θ + π/4)β 1 1 0 1 −α + {square root over (2)} cos(θ + 3π/4)β−α + {square root over (2)} sin(θ + 3π/4)β 0 1 1 1 −α + {square rootover (2)} cos(θ + 7π/4)β −α + {square root over (2)} sin(θ + 7π/4)β 0 10 1 −α + {square root over (2)} cos(θ + 5π/4)β −α + {square root over(2)} sin(θ + 5π/4)β

Referring to Table 3, each column defines the symbol position for eachfour (4) bits, s₃, s₂, s₁, s₀. Here, the position of each symbol isgiven in a two-dimensional signal space (m₁, m_(Q)). This means thateach symbol can be represented by S(t)=└M₁cos(2πf₀t+φ₀)+M_(Q)*sin(2πf₀t+φ₀)┘φ(t). Simply put, the complexmodulation symbol S=(m₁, m_(Q)) for each [s₃, s₂, s₁, s₀] is specifiedin S(t)=└M₁ cos(2πf₀t+φ₀)+M_(Q)*sin(2πf₀t+φ₀)┘φ(t).

Here, cos(2πf₀t+φ₀) and sin(2πf₀t+φ₀) denote the carrier signal withinitial phase φ₀ and carrier frequency f₀. Moreover, φ(t) denotes thepulse-shaping, the shape of a transmit symbol.

In the above definition of S(t), except the m₁ and m_(Q) value, otherparameters can usually either be shared between the transmitter and thereceiver or be detected by the receiver itself. For correctlydemodulating S(t), it is necessary to define and share the possiblevalue information of m₁ and m_(Q).

The possible value of m₁(k) and m_(Q)(k), which denote the m₁ and m_(Q)value for the k^(th) symbol, are given in Table 1. It shows forrepresenting each group inputs bits s₃, s₂, s₁, s₀ the symbol shall bemodulated by corresponding parameters shown in the table.

FIG. 19 is an exemplary diagram illustrating the signal constellation ofthe layered modulator with respect to 16 QAM/QPSK hierarchicalmodulation. Referring to another 16 QAM/QPSK hierarchical modulation,which means 16 QAM base layer and QPSK enhancement layer, eachmodulation symbol contains six (6) bits—s₅, s₄, s₃, s₂, s₁, s₀. The four(4) MSBs, s₅, s₄, s₃ and s₂, come from the base layer, and the two (2)LSBs, s₁ and s₀, come from the enhancement layer.

Given energy ratio r between the base layer and enhancement layer,$\alpha = {{\sqrt{\frac{r}{2\left( {1 + r} \right)}}\quad{and}\quad\beta} = \sqrt{\frac{1}{2\left( {1 + r} \right)}}}$can be defined such that 2(α²+β²)=1. Here, α denotes the amplitude ofthe base layer, and β denotes the amplitude of enhancement layer.Moreover, 2(α²+β²)=1 is a constraint which is also referred to as powerconstraint and more accurately referred to as normalization.

Table 4 illustrates a layered modulation table with 16 QAM base layerand QPSK enhancement layer. TABLE 4 Modulator Input Bits ModulationSymbols s₅ s₄ s₃ s₂ s₁ s₀ m_(I)(k) m_(Q)(k) 0 0 0 0 0 0 3α + {squareroot over (2)} cos(θ + π/4)β 3α + {square root over (2)} sin(θ + π/4)β 00 0 1 0 0 3α + {square root over (2)} cos(θ + 3π/4)β 3α + {square rootover (2)} sin(θ + 3π/4)β 1 0 0 0 0 0 3α + {square root over (2)} cos(θ +7π/4)β 3α + {square root over (2)} sin(θ + 7π/4)β 1 0 0 1 0 0 3α +{square root over (2)} cos(θ + 5π/4)β 3α + {square root over (2)}sin(θ + 5π/4)β 0 0 0 1 1 0 α + {square root over (2)} cos(θ + π/4)β 3α +{square root over (2)} sin(θ + π/4)β 0 0 0 0 1 0 α + {square root over(2)} cos(θ + 3π/4)β 3α + {square root over (2)} sin(θ + 3π/4)β 1 0 0 1 10 α + {square root over (2)} cos(θ + 7π/4)β 3α + {square root over (2)}sin(θ + 7π/4)β 1 0 0 0 1 0 α + {square root over (2)} cos(θ + 5π/4)β3α + {square root over (2)} sin(θ + 5π/4)β 0 0 0 1 0 1 −3α + {squareroot over (2)} cos(θ + π/4)β 3α + {square root over (2)} sin(θ + π/4)β 00 0 0 0 1 −3α + {square root over (2)} cos(θ + 3π/4)β 3α + {square rootover (2)} sin(θ + 3π/4)β 1 0 0 1 0 1 −3α + {square root over (2)}cos(θ + 7π/4)β 3α + {square root over (2)} sin(θ + 7π/4)β 1 0 0 0 0 1−3α + {square root over (2)} cos(θ + 5π/4)β 3α + {square root over (2)}sin(θ + 5π/4)β 0 0 0 0 1 1 −α + {square root over (2)} cos(θ + π/4)β3α + {square root over (2)} sin(θ + π/4)β 0 0 0 1 1 1 −α + {square rootover (2)} cos(θ + 3π/4)β 3α + {square root over (2)} sin(θ + 3π/4)β 1 00 0 1 1 −α + {square root over (2)} cos(θ + 7π/4)β 3α + {square rootover (2)} sin(θ + 7π/4)β 1 0 0 1 1 1 −α + {square root over (2)} cos(θ +5π/4)β 3α + {square root over (2)} sin(θ + 5π/4)β 1 1 0 0 0 0 3α +{square root over (2)} cos(θ + π/4)β α + {square root over (2)} sin(θ +π/4)β 1 1 0 1 0 0 3α + {square root over (2)} cos(θ + 3π/4)β α + {squareroot over (2)} sin(θ + 3π/4)β 0 1 0 0 0 0 3α + {square root over (2)}cos(θ + 7π/4)β α + {square root over (2)} sin(θ + 7π/4)β 0 1 0 1 0 03α + {square root over (2)} cos(θ + 5π/4)β α + {square root over (2)}sin(θ + 5π/4)β 1 1 0 1 1 0 α + {square root over (2)} cos(θ + π/4)β α +{square root over (2)} sin(θ + π/4)β 1 1 0 0 1 0 α + {square root over(2)} cos(θ + 3π/4)β α + {square root over (2)} sin(θ + 3π/4)β 0 1 0 1 10 α + {square root over (2)} cos(θ + 7π/4)β α + {square root over (2)}sin(θ + 7π/4)β 0 1 0 0 1 0 α + {square root over (2)} cos(θ + 5π/4)β α +{square root over (2)} sin(θ + 5π/4)β 1 1 0 1 0 1 −3α + {square rootover (2)} cos(θ + π/4)β α + {square root over (2)} sin(θ + π/4)β 1 1 0 00 1 −3α + {square root over (2)} cos(θ + 3π/4)β α + {square root over(2)} sin(θ + 3π/4)β 0 1 0 1 0 1 −3α + {square root over (2)} cos(θ +7π/4)β α + {square root over (2)} sin(θ + 7π/4)β 0 1 0 0 0 1 −3α +{square root over (2)} cos(θ + 5π/4)β α + {square root over (2)} sin(θ +5π/4)β 1 1 0 0 1 1 −α + {square root over (2)} cos(θ + π/4)β α + {squareroot over (2)} sin(θ + π/4)β 1 1 0 1 1 1 −α + {square root over (2)}cos(θ + 3π/4)β α + {square root over (2)} sin(θ + 3π/4)β 0 1 0 0 1 1−α + {square root over (2)} cos(θ + 7π/4)β α + {square root over (2)}sin(θ + 7π/4)β 0 1 0 1 1 1 −α + {square root over (2)} cos(θ + 5π/4)βα + {square root over (2)} sin(θ + 5π/4)β 1 0 1 0 0 0 3α + {square rootover (2)} cos(θ + π/4)β −3α + {square root over (2)} sin(θ + π/4)β 1 0 11 0 0 3α + {square root over (2)} cos(θ + 3π/4)β −3α + {square root over(2)} sin(θ + 3π/4)β 0 0 1 0 0 0 3α + {square root over (2)} cos(θ +7π/4)β −3α + {square root over (2)} sin(θ + 7π/4)β 0 0 1 1 0 0 3α +{square root over (2)} cos(θ + 5π/4)β −3α + {square root over (2)}sin(θ + 5π/4)β 1 0 1 1 1 0 α + {square root over (2)} cos(θ + π/4)β−3α + {square root over (2)} sin(θ + π/4)β 1 0 1 0 1 0 α + {square rootover (2)} cos(θ + 3π/4)β −3α + {square root over (2)} sin(θ + 3π/4)β 0 01 1 1 0 α + {square root over (2)} cos(θ + 7π/4)β −3α + {square rootover (2)} sin(θ + 7π/4)β 0 0 1 0 1 0 α + {square root over (2)} cos(θ +5π/4)β −3α + {square root over (2)} sin(θ + 5π/4)β 1 0 1 1 0 1 −3α +{square root over (2)} cos(θ + π/4)β −3α + {square root over (2)}sin(θ + π/4)β 1 0 1 0 0 1 −3α + {square root over (2)} cos(θ + 3π/4)β−3α + {square root over (2)} sin(θ + 3π/4)β 0 0 1 1 0 1 −3α + {squareroot over (2)} cos(θ + 7π/4)β −3α + {square root over (2)} sin(θ +7π/4)β 0 0 1 0 0 1 −3α + {square root over (2)} cos(θ + 5π/4)β −3α +{square root over (2)} sin(θ + 5π/4)β 1 0 1 0 1 1 −α + {square root over(2)} cos(θ + π/4)β −3α + {square root over (2)} sin(θ + π/4)β 1 0 1 1 11 −α + {square root over (2)} cos(θ + 3π/4)β −3α + {square root over(2)} sin(θ + 3π/4)β 0 0 1 0 1 1 −α + {square root over (2)} cos(θ +7π/4)β −3α + {square root over (2)} sin(θ + 7π/4)β 0 0 1 1 1 1 −α +{square root over (2)} cos(θ + 5π/4)β −3α + {square root over (2)}sin(θ + 5π/4)β 0 1 1 0 0 0 3α + {square root over (2)} cos(θ + π/4)β−α + {square root over (2)} sin(θ + π/4)β 0 1 1 1 0 0 3α + {square rootover (2)} cos(θ + 3π/4)β −α + {square root over (2)} sin(θ + 3π/4)β 1 11 0 0 0 3α + {square root over (2)} cos(θ + 7π/4)β −α + {square rootover (2)} sin(θ + 7π/4)β 1 1 1 1 0 0 3α + {square root over (2)} cos(θ +5π/4)β −α + {square root over (2)} sin(θ + 5π/4)β 0 1 1 1 1 0 α +{square root over (2)} cos(θ + π/4)β −α + {square root over (2)} sin(θ +π/4)β 0 1 1 0 1 0 α + {square root over (2)} cos(θ + 3π/4)β −α + {squareroot over (2)} sin(θ + 3π/4)β 1 1 1 1 1 0 α + {square root over (2)}cos(θ + 7π/4)β −α + {square root over (2)} sin(θ + 7π/4)β 1 1 1 0 1 0α + {square root over (2)} cos(θ + 5π/4)β −α + {square root over (2)}sin(θ + 5π/4)β 0 1 1 1 0 1 −3α + {square root over (2)} cos(θ + π/4)β−α + {square root over (2)} sin(θ + π/4)β 0 1 1 0 0 1 −3α + {square rootover (2)} cos(θ + 3π/4)β −α + {square root over (2)} sin(θ + 3π/4)β 1 11 1 0 1 −3α + {square root over (2)} cos(θ + 7π/4)β −α + {square rootover (2)} sin(θ + 7π/4)β 1 1 1 0 0 1 −3α + {square root over (2)}cos(θ + 5π/4)β −α + {square root over (2)} sin(θ + 5π/4)β 0 1 1 0 1 1−α + {square root over (2)} cos(θ + π/4)β −α + {square root over (2)}sin(θ + π/4)β 0 1 1 1 1 1 −α + {square root over (2)} cos(θ + 3π/4)β−α + {square root over (2)} sin(θ + 3π/4)β 1 1 1 0 1 1 −α + {square rootover (2)} cos(θ + 7π/4)β −α + {square root over (2)} sin(θ + 7π/4)β 1 11 1 1 1 −α + {square root over (2)} cos(θ + 5π/4)β −α + {square rootover (2)} sin(θ + 5π/4)β

Referring to Table 4, each column defines the symbol position for eachsix (6) bits, s₅, s₄, s₃, s₂, s₁, s₀. Here, the position of each symbolis given in a two-dimensional signal space (m₁, m_(Q)). This means thateach symbol can be represented by S(t)=└M₁cos(2πf₀t+φ₀)+M_(Q)*sin(2πf₀t+φ₀)┘φ(t). Simply put, the complexmodulation symbol S=(m₁, m_(Q)) for each [s₅, s₄, s₃, s₂, s₁, s₀] isspecified in S(t)=└M₁ cos(2πf₀t+φ₀)+M_(Q)*sin(2πf₀t+φ₀)┘φ(t).

Here, w₀ denotes carrier frequency, π₀ denotes an initial phase of thecarrier, and φ(t) denotes the symbol shaping or pulse shaping wave.Here, cos(2πf₀t+φ₀) and sin(2πf₀t+φ₀) denote the carrier signal withinitial phase φ₀ and carrier frequency f₀. Moreover, φ(t) denotes thepulse-shaping, the shape of a transmit symbol.

In the above definition of S(t), except the m₁ and m_(Q) value, otherparameters can usually either be shared between the transmitter and thereceiver or be detected by the receiver itself. For correctlydemodulating S(t), it is necessary to define and share the possiblevalue information of m₁ and m_(Q).

The possible value of m₁(k) and m_(Q)(k), which denote the m₁ and m_(Q)value for the k^(th) symbol, are given in Table 1. It shows forrepresenting each group inputs bits s₅, s₄, s₃, s₂, s₁, s₀ the symbolshall be modulated by corresponding parameters shown in the table.

With respect to the definitions of m₁ and m_(Q) in Table 1-4, inaddition to the contents, the show that the rotation angle θ also needsto be shared along with those tables between transmitter and receiver.Table 5 can be used to address this problem regarding how the receiverand transmitter share the rotation angle information.

To this end, Table 5 can be used which defines and/or maps four (4) bitsto a rotation angle. If this table is known by the receiver beforehand,then the transmitter only needs to sent four (4) bits to receiver toindicate to the receiver the initial rotation angle for demodulatingnext rotated layered modulated symbols. This table is an example ofquantizing the rotation angle θ with four (4) bits and uniformquantization. It is possible to quantize the rotation angle θ with othernumber of bits and different quantization rule for different accuracy.

More specifically, this table is either shared beforehand by thetransmitter and receiver (e.g., access network and access terminal),downloaded to the receiver (e.g., access terminal) over the air, or onlyused by the transmitter (e.g., access network) when the hierarchicalmodulation is enabled. The default rotation word for hierarchicalmodulation is 0000, which corresponds to 0.0.

Further, this table can be used by the receiver for demodulating therotated layered modulation, Compared with the regular or un-rotatedlayered modulation, the initial rotation angle is essentially zero (0).This information of initial rotation angle of zero (0) indicates animplicit consensus between the transmitter and the receiver. However,for rotated layered modulation, this information may not be implicitlyshared between the transmitter and/or the receiver. In other words, amechanism to send or indicate this initial rotation angle to thereceiver is necessary. TABLE 5 Mapped Bits for Rotation Angle (degree)Index Angle Rotating Unit: degree Unit: radian 0 0000 0.0 0.0 1 00012.81 0.04909 2 0011 5.63 0.09817 3 0010 8.44 0.1473 4 0110 11.25 0.19635 0111 14.06 0.2454 6 0101 16.88 0.2945 7 0100 19.69 0.3436 8 1100 22.500.3927 9 1101 25.31 0.4418 10 1111 28.13 0.4909 11 1110 30.94 0.5400 121010 33.75 0.5890 13 1011 36.56 0.6381 14 1001 39.38 0.6872 15 100042.19 0.7363

In a further application of the layered or superposition modulation forBCMCS, layered modulation can be a superposition of any two modulationschemes. In BCMCS, a QPSK enhancement layer is superposed on a base QPSKor 16-QAM layer to obtain the resultant signal constellation. The energyratio r is the power ratio between the base layer and the enhancement.Furthermore, the enhancement layer is rotated by the angle incounter-clockwise direction.

FIG. 20 is an exemplary diagram illustrating signal constellation forlayered modulation with QPSK base layer and QPSK enhancement layer.Referring to FIG. 20, each modulation symbol contains four (4) bits,namely, s₃, s₂, s₁, s₀. Here, there are two (2) MSBs which are s₃ ands₁, and two (2) LSBs which are s₂ and s₀. The two (2) MSBs are from thebase layer and the two LSBs come from the enhancement layer

Given energy ratio r between the base layer and enhancement layer,$\alpha = {{\sqrt{\frac{r}{2\left( {1 + r} \right)}}\quad{and}\quad\beta} = \sqrt{\frac{1}{2\left( {1 + r} \right)}}}$can be defined such that 2(α²+β²)=1. Here, α denotes the amplitude ofthe base layer, and α denotes the amplitude of enhancement layer.Moreover, 2(α²'β²)=1 is a constraint which is also referred to as powerconstraint and more accurately referred to as normalization.

Table 6 illustrates a layered modulation table with QPSK base layer andQPSK enhancement layer. TABLE 6 Modulator Input Bits Modulation Symbolss₃ s₂ s₁ s₀ m_(I)(k) m_(Q)(k) 0 0 0 0 α + {square root over (2)} cos(θ +π/4)β α + {square root over (2)} sin(θ + π/4)β 0 0 1 0 α + {square rootover (2)} cos(θ + 3π/4)β α + {square root over (2)} sin(θ + 3π/4)β 1 0 00 α + {square root over (2)} cos(θ + 7π/4)β α + {square root over (2)}sin(θ + 7π/4)β 1 0 1 0 α + {square root over (2)} cos(θ + 5π/4)β α +{square root over (2)} sin(θ + 5π/4)β 0 0 1 1 −α + {square root over(2)} cos(θ + π/4)β α + {square root over (2)} sin(θ + π/4)β 0 0 0 1 −α +{square root over (2)} cos(θ + 3π/4)β α + {square root over (2)} sin(θ +3π/4)β 1 0 1 1 −α + {square root over (2)} cos(θ + 7π/4)β α + {squareroot over (2)} sin(θ + 7π/4)β 1 0 0 1 −α + {square root over (2)}cos(θ + 5π/4)β α + {square root over (2)} sin(θ + 5π/4)β 1 1 0 0 α +{square root over (2)} cos(θ + π/4)β −α + {square root over (2)} sin(θ +π/4)β 1 1 1 0 α + {square root over (2)} cos(θ + 3π/4)β −α + {squareroot over (2)} sin(θ + 3π/4)β 0 1 0 0 α + {square root over (2)} cos(θ +7π/4)β −α + {square root over (2)} sin(θ + 7π/4)β 0 1 1 0 α + {squareroot over (2)} cos(θ + 5π/4)β −α + {square root over (2)} sin(θ + 5π/4)β1 1 1 1 −α + {square root over (2)} cos(θ + π/4)β −α + {square root over(2)} sin(θ + π/4)β 1 1 0 1 −α + {square root over (2)} cos(θ + 3π/4)β−α + {square root over (2)} sin(θ + 3π/4)β 0 1 1 1 −α + {square rootover (2)} cos(θ + 7π/4)β −α + {square root over (2)} sin(θ + 7π/4)β 0 10 1 −α + {square root over (2)} cos(θ + 5π/4)β −α + {square root over(2)} sin(θ + 5π/4)β

Referring to Table 6, each column defines the symbol position for eachfour (4) bits, s₃, s₂, s₁, s₀. Here, the position of each symbol isgiven in a two-dimensional signal space (m₁, m_(Q)). This means thateach symbol can be represented by S(t)=M₁cos(2πf₀t+φ₀)+M_(Q)*sin(2πf₀t+φ₀)┘φ(t). Simply put, the complexmodulation symbol S=(m₁, m_(Q)) for each [s₃, s₂, s₁, s₀] is specifiedin S(t)=└M₁ cos(2πf₀t+φ₀)+M_(Q)*sin(2πf₀t+φ₀)┘φ(t).

Here, cos(2πf₀t+φ₀) and sin(2πf₀t+φ₀) denote the carrier signal withinitial phase φ₀ and carrier frequency f₀. Moreover, φ(t) denotes thepulse-shaping, the shape of a transmit symbol.

In the above definition of S(t), except the m₁ and m_(Q) value, otherparameters can usually either be shared between the transmitter and thereceiver or be detected by the receiver itself. For correctlydemodulating S(t), it is necessary to define and share the possiblevalue information of m₁ and m_(Q).

The possible value of m₁(k) and m_(Q)(k), which denote the m₁ and m_(Q)value for the k^(th) symbol, are given in Table 1. It shows forrepresenting each group inputs bits s₃, s₂, s₁, s₀ the symbol shall bemodulated by corresponding parameters shown in the table.

FIG. 21 is an exemplary diagram illustrating the signal constellation ofthe layered modulator with respect to 16 QAM/QPSK hierarchicalmodulation. Referring to another 16 QAM/QPSK hierarchical modulation,which means 16 QAM base layer and QPSK enhancement layer, eachmodulation symbol contains six (6) bits—s₅, s₄, s₃, s₂, s₁, s₀. The four(4) MSBs, s₄, s₃, s₁ and s₀, come from the base layer, and the two (2)LSBs, s₅ and s₂, come from the enhancement layer.

Given energy ratio r between the base layer and enhancement layer,$\alpha = {{\sqrt{\frac{r}{2\left( {1 + r} \right)}}\quad{and}\quad\beta} = \sqrt{\frac{1}{2\left( {1 + r} \right)}}}$can be defined such that 2(α²+β²)=1. Here, α denotes the amplitude ofthe base layer, and β denotes the amplitude of enhancement layer.Moreover, 2(α²+β²)=1 is a constraint which is also referred to as powerconstraint and more accurately referred to as normalization.

Table 7 illustrates a layered modulation table with 16QAM base layer andQPSK enhancement layer. TABLE 7 Modulator Input Bits Modulation Symbolss₅ s₄ s₃ s₂ s₁ s₀ m_(I)(k) m_(Q)(k) 0 0 0 0 0 0 3α + {square root over(2)} cos(θ + π/4)β 3α + {square root over (2)} sin(θ + π/4)β 0 0 0 1 0 03α + {square root over (2)} cos(θ + 3π/4)β 3α + {square root over (2)}sin(θ + 3π/4)β 1 0 0 0 0 0 3α + {square root over (2)} cos(θ + 7π/4)β3α + {square root over (2)} sin(θ + 7π/4)β 1 0 0 1 0 0 3α + {square rootover (2)} cos(θ + 5π/4)β 3α + {square root over (2)} sin(θ + 5π/4)β 0 00 1 0 1 α + {square root over (2)} cos(θ + π/4)β 3α + {square root over(2)} sin(θ + π/4)β 0 0 0 0 0 1 α + {square root over (2)} cos(θ + 3π/4)β3α + {square root over (2)} sin(θ + 3π/4)β 1 0 0 1 0 1 α + {square rootover (2)} cos(θ + 7π/4)β 3α + {square root over (2)} sin(θ + 7π/4)β 1 00 0 0 1 α + {square root over (2)} cos(θ + 5π/4)β 3α + {square root over(2)} sin(θ + 5π/4)β 0 0 0 1 1 0 −3α + {square root over (2)} cos(θ +π/4)β 3α + {square root over (2)} sin(θ + π/4)β 0 0 0 0 1 0 −3α +{square root over (2)} cos(θ + 3π/4)β 3α + {square root over (2)}sin(θ + 3π/4)β 1 0 0 1 1 0 −3α + {square root over (2)} cos(θ + 7π/4)β3α + {square root over (2)} sin(θ + 7π/4)β 1 0 0 0 1 0 −3α + {squareroot over (2)} cos(θ + 5π/4)β 3α + {square root over (2)} sin(θ + 5π/4)β0 0 0 0 1 1 −α + {square root over (2)} cos(θ + π/4)β 3α + {square rootover (2)} sin(θ + π/4)β 0 0 0 1 1 1 −α + {square root over (2)} cos(θ +3π/4)β 3α + {square root over (2)} sin(θ + 3π/4)β 1 0 0 0 1 1 −α +{square root over (2)} cos(θ + 7π/4)β 3α + {square root over (2)}sin(θ + 7π/4)β 1 0 0 1 1 1 −α + {square root over (2)} cos(θ + 5π/4)β3α + {square root over (2)} sin(θ + 5π/4)β 1 0 1 0 0 0 3α + {square rootover (2)} cos(θ + π/4)β α + {square root over (2)} sin(θ + π/4)β 1 0 1 10 0 3α + {square root over (2)} cos(θ + 3π/4)β α + {square root over(2)} sin(θ + 3π/4)β 0 0 1 0 0 0 3α + {square root over (2)} cos(θ +7π/4)β α + {square root over (2)} sin(θ + 7π/4)β 0 0 1 1 0 0 3α +{square root over (2)} cos(θ + 5π/4)β α + {square root over (2)} sin(θ +5π/4)β 1 0 1 1 0 1 α + {square root over (2)} cos(θ + π/4)β α + {squareroot over (2)} sin(θ + π/4)β 1 0 1 0 0 1 α + {square root over (2)}cos(θ + 3π/4)β α + {square root over (2)} sin(θ + 3π/4)β 0 0 1 1 0 1 α +{square root over (2)} cos(θ + 7π/4)β α + {square root over (2)} sin(θ +7π/4)β 0 0 1 0 0 1 α + {square root over (2)} cos(θ + 5π/4)β α + {squareroot over (2)} sin(θ + 5π/4)β 1 0 1 1 1 0 −3α + {square root over (2)}cos(θ + π/4)β α + {square root over (2)} sin(θ + π/4)β 1 0 1 0 1 0 −3α +{square root over (2)} cos(θ + 3π/4)β α + {square root over (2)} sin(θ +3π/4)β 1 0 1 1 1 0 −3α + {square root over (2)} cos(θ + 7π/4)β α +{square root over (2)} sin(θ + 7π/4)β 0 0 1 0 1 0 −3α + {square rootover (2)} cos(θ + 5π/4)β α + {square root over (2)} sin(θ + 5π/4)β 1 0 10 1 1 −α + {square root over (2)} cos(θ + π/4)β α + {square root over(2)} sin(θ + π/4)β 1 0 1 1 1 1 −α + {square root over (2)} cos(θ +3π/4)β α + {square root over (2)} sin(θ + 3π/4)β 0 0 1 0 1 1 −α +{square root over (2)} cos(θ + 7π/4)β α + {square root over (2)} sin(θ +7π/4)β 0 0 1 1 1 1 −α + {square root over (2)} cos(θ + 5π/4)β α +{square root over (2)} sin(θ + 5π/4)β 1 1 0 0 0 0 3α + {square root over(2)} cos(θ + π/4)β −3α + {square root over (2)} sin(θ + π/4)β 1 1 0 1 00 3α + {square root over (2)} cos(θ + 3π/4)β −3α + {square root over(2)} sin(θ + 3π/4)β 0 1 0 0 0 0 3α + {square root over (2)} cos(θ +7π/4)β −3α + {square root over (2)} sin(θ + 7π/4)β 0 1 0 1 0 0 3α +{square root over (2)} cos(θ + 5π/4)β −3α + {square root over (2)}sin(θ + 5π/4)β 1 1 0 1 0 1 α + {square root over (2)} cos(θ + π/4)β−3α + {square root over (2)} sin(θ + π/4)β 1 1 0 0 0 1 α + {square rootover (2)} cos(θ + 3π/4)β −3α + {square root over (2)} sin(θ + 3π/4)β 0 10 1 0 1 α + {square root over (2)} cos(θ + 7π/4)β −3α + {square rootover (2)} sin(θ + 7π/4)β 0 1 0 0 0 1 α + {square root over (2)} cos(θ +5π/4)β −3α + {square root over (2)} sin(θ + 5π/4)β 1 1 0 1 1 0 −3α +{square root over (2)} cos(θ + π/4)β −3α + {square root over (2)}sin(θ + π/4)β 1 1 0 0 1 0 −3α + {square root over (2)} cos(θ + 3π/4)β−3α + {square root over (2)} sin(θ + 3π/4)β 0 1 0 1 1 0 −3α + {squareroot over (2)} cos(θ + 7π/4)β −3α + {square root over (2)} sin(θ +7π/4)β 0 1 0 0 1 0 −3α + {square root over (2)} cos(θ + 5π/4)β −3α +{square root over (2)} sin(θ + 5π/4)β 1 1 0 0 1 1 −α + {square root over(2)} cos(θ + π/4)β −3α + {square root over (2)} sin(θ + π/4)β 1 1 0 1 11 −α + {square root over (2)} cos(θ + 3π/4)β −3α + {square root over(2)} sin(θ + 3π/4)β 0 1 0 0 1 1 −α + {square root over (2)} cos(θ +7π/4)β −3α + {square root over (2)} sin(θ + 7π/4)β 0 1 0 1 1 1 −α +{square root over (2)} cos(θ + 5π/4)β −3α + {square root over (2)}sin(θ + 5π/4)β 0 1 1 0 0 0 3α + {square root over (2)} cos(θ + π/4)β−α + {square root over (2)} sin(θ + π/4)β 0 1 1 1 0 0 3α + {square rootover (2)} cos(θ + 3π/4)β −α + {square root over (2)} sin(θ + 3π/4)β 1 11 0 0 0 3α + {square root over (2)} cos(θ + 7π/4)β −α + {square rootover (2)} sin(θ + 7π/4)β 1 1 1 1 0 0 3α + {square root over (2)} cos(θ +5π/4)β −α + {square root over (2)} sin(θ + 5π/4)β 0 1 1 1 0 1 α +{square root over (2)} cos(θ + π/4)β −α + {square root over (2)} sin(θ +π/4)β 0 1 1 0 0 1 α + {square root over (2)} cos(θ + 3π/4)β −α + {squareroot over (2)} sin(θ + 3π/4)β 1 1 1 1 0 1 α + {square root over (2)}cos(θ + 7π/4)β −α + {square root over (2)} sin(θ + 7π/4)β 1 1 1 0 0 1α + {square root over (2)} cos(θ + 5π/4)β −α + {square root over (2)}sin(θ + 5π/4)β 0 1 1 1 1 0 −3α + {square root over (2)} cos(θ + π/4)β−α + {square root over (2)} sin(θ + π/4)β 0 1 1 0 1 0 −3α + {square rootover (2)} cos(θ + 3π/4)β −α + {square root over (2)} sin(θ + 3π/4)β 1 11 1 1 0 −3α + {square root over (2)} cos(θ + 7π/4)β −α + {square rootover (2)} sin(θ + 7π/4)β 1 1 1 0 1 0 −3α + {square root over (2)}cos(θ + 5π/4)β −α + {square root over (2)} sin(θ + 5π/4)β 0 1 1 0 1 1−α + {square root over (2)} cos(θ + π/4)β −α + {square root over (2)}sin(θ + π/4)β 0 1 1 1 1 1 −α + {square root over (2)} cos(θ + 3π/4)β−α + {square root over (2)} sin(θ + 3π/4)β 1 1 1 0 1 1 −α + {square rootover (2)} cos(θ + 7π/4)β −α + {square root over (2)} sin(θ + 7π/4)β 1 11 1 1 1 −α + {square root over (2)} cos(θ + 5π/4)β −α + {square rootover (2)} sin(θ + 5π/4)β

Referring to Table 4, each column defines the symbol position for eachsix (6) bits, s₅, s₄, s₃, s₁, s₀. Here, the position of each symbol isgiven in a two-dimensional signal space (m₁, m_(Q)). This means thateach symbol can be represented by S(t)=└M₁cos(2πf₀t+φ₀)+M_(Q)*sin(2πf₀t+φ₀)┘φ(t). Simply put, the complexmodulation symbol S=(m₁, m_(Q)) for each [s₅, s₄, s₃, s₂, s₁, s₀] isspecified in S(t)=└M₁ cos(2πf₀t+φ₀)+M_(Q)*sin(2πf₀t+φ₀)┘φ(t).

Here, w₀ denotes carrier frequency, π₀ denotes an initial phase of thecarrier, and φ(t) denotes the symbol shaping or pulse shaping wave.Here, cos(2πf₀t+φ₀) and sin(2πf₀t+φ₀) denote the carrier signal withinitial phase φ₀ and carrier frequency f₀. Moreover, φ(t) denotes thepulse-shaping, the shape of a transmit symbol.

In the above definition of S(t), except the m₁ and m_(Q) value, otherparameters can usually either be shared between the transmitter and thereceiver or be detected by the receiver itself. For correctlydemodulating S(t), it is necessary to define and share the possiblevalue information of m₁ and m_(Q).

The possible value of m₁(k) and m_(Q)(k), which denote the m₁ and m_(Q)value for the k^(th) symbol, are given in Table 1. It shows forrepresenting each group inputs bits s₅, s₄, s₃, s₂, s₁, s₀ the symbolshall be modulated by corresponding parameters shown in the table.

However, the Euclid distance profile can change when the enhancementlayer signal constellation is rotated and the power-splitting ratio ischanged. This means the original Gray mapping in Error! Reference sourcenot found.1, for example, may not always be optimal. In this case, itmay be necessary to perform bits-to-symbols remapping based on eachEuclidean distance file instance. FIG. 22 is an exemplary diagramillustrating Gray mapping for rotated QPSK/QPSK hierarchical modulation.

The HER performance of a signal constellation can be dominated by symbolpairs with minimum Euclidean distance, especially when SNR is high.Therefore it is interesting to find optimal bits-to-symbol mappingrules, in which the codes for the closest two signals have minimumdifference.

In general, Gray mapping in two-dimensional signals worked with channelcoding can be accepted as optimal for minimizing HER for equally likelysignals. Gray mapping for regular hierarchical signal constellations isshown in FIG. 21, where the codes for the closest two signals aredifferent in only one bit. However, this kind of Euclidean distanceprofile may not be fixed in hierarchical modulation. An example of theminimum Euclidean distance of 16 QAM/QPSK hierarchical modulation withdifferent rotation angles is shown in FIG. 23.

FIG. 23 is an exemplary diagram illustrating an enhanced QPSK/QPSKhierarchical modulation. Referring to FIG. 23, the base layered ismodulated with QPSK and the enhancement layer is modulated with rotatedQPSK. If the hierarchical modulation is applied, a new QPSK/QPSKhierarchical modulation can be attained as shown in this figure.

Further, the inter-layer Euclidean distance may become shortest when thepower splitting ratio increases in a two-layer hierarchical modulation.This can occur if the enhancement layer is rotated. In order to minimizeBER when Euclidean distance profile is changed in hierarchicalmodulation, the bits-to-symbol mapping can be re-done or performedagain, as shown in FIGS. 24 and 25.

FIG. 24 is an exemplary diagram illustrating a new QPSK/QPSKhierarchical modulation. Moreover, FIG. 25 is another exemplary diagramillustrating a new QPSK/QPSK hierarchical modulation.

In view of the discussions of above, a new bit-to-symbol generationstructure can be introduced. According to the conventional structure, asymbol mapping mode selection was not available. FIG. 26 is an exemplarydiagram illustrating a new bit-to-symbol block. Here, the symbol mappingmode can be selected when the bits-to-symbol mapping is performed. Morespecifically, a new symbol mapping mode selection block can be added forcontrolling and/or selecting bits-to-symbol mapping rule based on thesignal constellation of hierarchical modulation and channel coding used.

It will be apparent to those skilled in the art that variousmodifications and variations can be made in the present inventionwithout departing from the spirit or scope of the inventions. Thus, itis intended that the present invention covers the modifications andvariations of this invention provided they come within the scope of theappended claims and their equivalents.

1. A method of allocating symbols in a wireless communication system,the method comprising: receiving at least one data stream from at leastone user; grouping the at least one data streams into at least onegroup, wherein each group is comprised of at least one data stream;preceding each group of data streams in multiple stages; and allocatingthe precoded symbols.
 2. The method of claim 1, wherein the each groupof data streams is precoded independently.
 3. The method of claim 2,wherein the each group of data stream is precoded independently usingindependent rotation matrix.
 4. The method of claim 1, wherein the eachgroup of data streams is precoded jointly.
 5. The method of claim 4,wherein the each group of data streams is precoded jointly using asingle rotation matrix.
 6. The method of claim 1, wherein the precodingin multiple stages include applying independent spreading matrix to eachgroup.
 7. The method of claim 1, wherein the preceding includes at leastone of phase adjustment or amplitude adjustment.
 8. The method of claim1, wherein the wireless communication system is any one of orthogonalfrequency division multiplexing (OFDM) system, orthogonal frequencydivision multiple access (OFDMA) system, multi-carrier code divisionmultiplexing (MC-CDM), or multi-carrier code division multiple access(MC-CDMA).
 9. The method of claim 1, further comprising modulating theallocated symbols using an inverse fast Fourier transform (IFFT) or aninverse discrete Fourier transform (IDFT).
 10. A method of performinghierarchical modulation signal constellation in a wireless communicationsystem, the method comprising allocating multiple symbols according to abits-to-symbol mapping rule representing different signal constellationpoints with different bits, wherein the mapping rule represents one (1)or less bit difference between closest two symbols.
 11. The method ofclaim 10, wherein the multiple symbols have different initial modulationphase.
 12. The method of claim 10, wherein the hierarchical modulationsignal constellation includes one base layer signal constellation and atleast one enhancement layer signal constellation.
 13. The method ofclaim 12, wherein the mapping rule applied to the enhancement layer isselected from a pool of all possible enhancement layer mapping ruleswhich is based on each base layer symbol position.
 14. The method ofclaim 10, wherein the hierarchical modulation signal constellationincludes one base layer signal constellation and at least oneenhancement layer signal constellation and the mapping rule representedby a bit-to-symbol mapping rule.
 15. The method of claim 10, furthercomprising multiplexing the bits for base layer symbol and the bits forenhancement layer symbol using interleaving or concatenating techniques.16. The method of claim 10, further comprising: grouping the symbols,each group having the same signal strength; and selecting the each groupfrom a pool of mapping rules according to the bits-to-symbol mappingrule applied to other groups.
 17. The method of claim 10, wherein themodulation schemes include phase shift keying (PSK), rotated-PSK,quadrature phase shift keying (QPSK), rotated-QPSK, 8-PSK, rotated8-PSK, 16 quadrature amplitude modulation (16 QAM), and rotated-16 QAM.18. The method of claim 10, wherein the bits-to-symbol mapping m/e isGray mapping rule.
 19. A method of transmitting more than one signal ina wireless communication system, the method comprising: allocatingmultiple symbols to a first signal constellation and to a secondconstellation, wherein the first signal constellation refers to baselayer signals and the second signal constellation refers to enhancementlayer signals; modulating the multiple symbols of the first signalconstellation and the second signal constellation; and transmitting themodulated symbols.
 20. The method of claim 19, wherein the base layersignals and the enhancement layer signals have initial modulation andtransmission phase that are the same.
 21. The method of claim 19,wherein the base layer signals and the enhancement layer signals haveinitial modulation and transmission phase that are different.
 22. Themethod of claim 19, wherein the base layer signals and the enhancementlayer signals have the same bits-to-symbol mapping rules.
 23. The methodof claim 19, wherein the base layer signals and the enhancement layersignals have different bits-to-symbol mapping rules.
 24. The method ofclaim 19, wherein the transmitted modulated symbols apply bits-to-symbolmapping rule where each enhancement layer signal constellation is basedon bits-to-symbol mapping rule for the base layer bits-to-symbol mappingrule and other enhancement bits-to-symbol mapping rule.